1. Technical Field
The present disclosure generally relates to switching power amplifier, and more particularly to single stage switching power amplifier which, among other advantages, achieve bidirectional energy flow.
2. Description of the Related Art
A conventional switching power amplifier, such as a class-D power amplifier, usually undergoes at least two stages—namely, the first stage of generating needed power supply (hereinafter referred to as “the power-supplying stage”) and the second stage of applying the generated power supply to amplify an input signal with one or more switching configurations (hereinafter referred to as “the power-amplifying stage”)—before producing an output signal across a load coupled to the output terminals of the power amplifier, with the output signal being an amplified replica of the input signal. At each stage, such a two-stage or multi-stage power amplifier incurs power losses, resulting in lowering the system efficiency. Thus, for a switching power amplifier, reducing the number of stages is one way to increase the system efficiency.
In increasing the system efficiency of a switching power amplifier, efforts have been made to combine the first “power-supplying” stage and the second “power-amplifying” stage into a single stage in delivering an amplified output signal. In particular, U.S. Patent Application Publication No. 2011/0299309 to Chen (hereinafter referred to as “the '309 application”) discloses a switching power amplifier (hereinafter referred to “the '309 amplifier”) where the “power-supplying” stage and the “power-amplifying” stage are combined into a single stage to increase the system efficiency. However, the '309 amplifier suffers from a number of design deficiencies, primarily due to its failure to adequately address new issues arising from combining the “power-supplying” stage and the “power-amplifying” stage into a single stage.
Referring to FIG. 1A, which shows a schematic circuit diagram of the '309 power amplifier, one design deficiency is that the '309 power amplifier requires a highly magnetically coupled pair of inductors L5 and L10 (as respective integral parts of a pair of low pass filters) across load 150 with a coupling coefficient better than 0.99, in order for the amplifier to work properly. FIGS. 1B and 1C illustrate examples of the required highly coupled pair of inductors.
To be more specific, diodes D131 and D134, the two diodes (on the secondary winding) coupled to inductor L5 via node A, are both biased in the reverse direction with respect to inductor L5. Thus, when MOSFET switch M1 (which is also coupled to node A) is turned off, diodes D131 and D134 block the current flowing in inductor L5. As a skilled artisan appreciates, if the current flowing in an inductor is blocked, a high voltage spike at the blocking end of the inductor will be caused, unless the inductor is tightly magnetically coupled with another inductor so as to induce a negative voltage on the magnetically coupled end of the other inductor to create a current flow circulation.
Thus, when the current flowing in inductor L5 is blocked by the reverse biasing of diodes D131 and D134, a high voltage spike at node A will occur (which may destroy switch M1), unless inductors L5 and L10 are tightly magnetically coupled so as to induce a negative voltage at node B (see FIGS. 1B and 1C) to create a current flow circulation through the body diode of MOSFET switch M2. Further, the coupling coefficient between inductors L5 and L10 must be better than 0.99 or the coupling would not be able to instantaneously induce the voltage of node B to go negative, which is a necessity to create the needed current flow circulation. The same situation is also applicable to inductor L10. Accordingly, without a tight magnetic coupling between inductor L5 and L10 with a coupling coefficient 0.99 or better, the '309 amplifier simply will not work properly.
As a skilled artisan also appreciates, although it may be possible to make a tight coupling of inductors L5 and L10 having a coupling coefficient of 0.99 or higher in a lab environment, it is practically impossible to make such a tight magnetic coupling between inductors in mass production. Even if such an extremely high level of coupling could be materialized in mass production, the coupling would inevitably cause a high-frequency ripple current—whose peak current can go as high as 100 amps—flowing through L5 and L10. The high-frequency ripple current can cause inductor core losses, since, with today's technology, very few (if not no) magnetic material (of which any inductor is made) can sustain such an extreme high-frequency ripple current.
Besides, the ripple current, if generated, would also flow through capacitors C1 and C2 (which are connected to L5 and L10 as integral parts of the pair of low pass filters, respectively). With today's technology, however, it is practically impossible to find capacitors that can handle this amount of ripple current. Furthermore, with the tightly coupled inductors L5 and L10, there also comes the leakage inductance thereof, which causes high-magnitude voltage spikes on both nodes A and B of the '309 amplifier. Such voltage spikes, even if controlled with a clamping circuit, will cause lots of energy wasted, thereby reducing the system efficiency. Thus, for the '309 amplifier, requiring a highly magnetically coupled pair of inductors L5 and L10 is a design deficiency.
The second design deficiency of the '309 power amplifier is that the '309 power amplifier can only drive a resistive type of load, but cannot drive a pure inductive type of load. In particular, under its design, the '309 amplifier does not provide storage as well as a return path for excess energy released (returned) from the inductive load. As a result, the '309 amplifier cannot drive a pure inductive type of load, such as a motor. This limitation severely restricts the usability of the '309 amplifier, and thus is also a design deficiency.
The third design deficiency of the '309 power amplifier is that its configuration in connection with MOSFET switches M1 and M2 cannot reliably ensure that only one of M1 and M2, but not both, is conducting at any moment. More specifically, the signals generated at the upper secondary winding are used as driver signals to drive switch M1 through diodes D142 and D143, both of which are biased in the reverse direction with respect to M1. As a skilled artisan appreciates, due to excessive charges accumulated on the driving gate of M1 resulting from the reverse-biasing of D142 and D143, M1 simply cannot be reliably turned off before or when M2 is turned on, resulting both switches being on at the same time for an extended length of time. This is a condition that can cause the secondary windings of the two switching transformers shorted and as a result render the '309 amplifier inoperable. The same situation also applies to switch M2. Thus, for the '309 amplifier, the present configuration in connection with switches M1 and M2 is yet another design deficiency.
The aforementioned design deficiencies of the '309 amplifier are primarily due to its failure to address issues arising from combining the aforementioned power-supplying stage and the power-amplifying stage into a single stage. These issues are usually not applicable to a conventional two-stage switching power amplifier.
Referring to FIG. 1D, which illustrates a conventional class D switching power amplifier 100, the power-amplifying, as shown, takes place on the secondary side of one or more transformers providing the required power supply. Typically, the power-amplifying circuit includes one set of four switches S101 to S104 connected to two low pass filters situated across load 150. As well known, each switch has an internal body-diode situated between its source and drain. Additionally, two filter capacitors C101 and C102 on the secondary side, which are integral parts of the power-supplying stage, are coupled to each of switches S101-S104 as well as coupled to positive and negative rails of a power supply (such as the secondary winding of a switching transformer).
In particular, the pair of inductors L101 and L102, to some extent, correspond to the pair of inductors L5 and L10 of the '309 amplifier, since both pairs of inductors are associated with a pair of low pass filters situated on both sides of load 150. Additionally, switches S102 and S104, to some extent, correspond to switches M1 and M2 of the '309 amplifier, since both pairs of switches are coupled to their respective pairs of low pass filters across load 150.
First, in a scenario similar to the one associated with the first design deficiency of the '309 amplifier—namely, the scenario where when S102 is turned off, the current flowing in inductor L101 is blocked by switch S101—the body diode D101 of switch S101 is conducted. As a result, the current of inductor L101 flows through body diode D101 to charge filter capacitor C101 and C102. As a skilled artisan appreciates, filter capacitors C101 and C102 work like an energy storage tank. As the inductor current keeps flowing through body diode D101 of power switch S101, the current keeps charging filter capacitors C101 and C102. As long as the capacitors are configured to have enough capacitance, excess energy can be stored therein, thus enabling the inductor current to continue to flow. This is different from the aforementioned situation of the '309 amplifier where the inductor current will stop flowing and cause high voltage spike at node A unless inductors L5 and L10 are tightly magnetically coupled. Therefore, unlike the '309 amplifier, conventional switching amplifier 100, in a similar situation, usually does not require that a tight magnetic coupling of inductors be present across load 150.
Next, in a scenario similar to the one associated with the second design deficiency of the '309 amplifier—namely, the scenario where there is considerable energy returned from a pure inductive load—capacitors C101 and C102 of conventional amplifier 100, as noted, provide storage as well as a current circulation path for excess energy returned from the pure inductive load. Thus, unlike the '309 amplifier, conventional switching amplifier 100 does not have an issue with regard to handling a pure inductive load.
In view of the differences in dealing with similar scenarios between the single-stage '309 amplifier and conventional two-stage switching amplifier 100, it should be apparent to a skilled artisan that those differences are mostly resulted from combining two stages into one stage. When the power-supplying stage and the power-amplifying stage are combined into one stage, capacitors C101 and C102—which, as shown in FIG. 1D, are parts of the power-supplying stage—may not be available in the newly combined single stage in a manner availed by a conventional switching amplifier. This is primarily because arrangements and availability with respect to switches on the secondary side of one or more switching transformers are changed with the stage-combining. As a result, new issues—such as not readily having a path for high-frequency inductor current when one switch is turned off, not readily having storage as well as a current circulation path for energy returned from a pure inductive load, and not readily having much flexibility in relative timings between switches—inevitably arise. As to the design of the '309 amplifier, these new issues are simply not adequately addressed, thus resulting in the aforementioned design deficiencies.
Therefore, in implementing a single-stage switch amplifier to increase system efficiency, there is a need to provide a new design distinct from and departing from that of the '309 amplifier in order to adequately new issues arising from combining two stages into one single stage while achieving the system efficiency advantage that comes with the stage-combining.